1. Field of the Invention
The present invention relates to a transimpedance amplifier, an optical communication module and an optical communication apparatus which includes the optical communication module, and more particularly to an optical communication apparatus such as a router or a server and a high sensitive transimpedance amplifier which is part of the optical communication apparatus and is effectively applicable to an optical communication module which performs optical communication using a laser diode and a photo diode.
2. Description of the Related Art
For example, JP-A-2011-66751 (patent literature 1) discloses a transimpedance amplifier which converts a current signal from a photo diode into a voltage signal and amplifies the voltage signal in a high-speed optical receiver circuit. To be more specific, patent literature discloses the transimpedance amplifier which is constituted of a pre-amplifier, a post-amplifier, an error amplifier, and an auto-threshold voltage control circuit, wherein a negative feedback path is formed by the post-amplifier, the error amplifier, and an automatic threshold voltage control circuit so that an offset voltage of the post-amplifier can be reduced by a gain of the negative feedback path.
Recently, along with a demand for a higher communication speed, the communication speed has been increased to 25 Gbps, 40 Gbps or the like from 10 Gbps. To cope with such a demand for a higher communication speed, the application of an optical communication apparatus corresponding to an optical fiber cable to a router device or a server device, for example, has been steadily progressing. The optical communication apparatus is usually designed on a premise that the optical communication apparatus is used for the long distance transmission at the order of kilometers between devices, and it is important to ensure the high speed property and reliability in such long distance transmission.
Among such optical communication apparatuses, there have been known many devices which have a relatively large size (for example, the order of several tens of centimeters to the order of meters). Inside of these apparatuses, usually, the communication is performed using an electrical signal. That is, the optical communication apparatus, for example, converts an optical signal inputted from the outside into an electrical signal, performs predetermined processing through short distance communication (for example, of the order of meters) in the inside of the apparatus using the electrical signal, and converts the electrical signal again into an optical signal, and outputs the optical signal to the outside. In such short distance communication, communication by an electrical signal is performed using a copper line cable or the like, for example. However, along with the demand for a higher communication speed, quality of a transmission waveform is remarkably deteriorated when the communication is performed using the copper line cable. To cope with such a situation, a demand for using optical communication also in such short distance communication in the inside of the apparatus has been increasing. In this case, in the optical communication, all internal signal-processings in a router or the like are performed using an electrical signal and hence, it is necessary to convert an optical signal into an electrical signal by an optical element. Accordingly, it is desirable to suppress electric power consumed by such a portion to a small value.
FIG. 1A shows an image view of the constitution of a router apparatus where an optical communication module is applied to communication between cards. In this constitution, optical communication is performed between LSI_LGi mounted on interface cards IFC[1], IFC[2] and switches LSI_LGs mounted on a switch card SWC by optical communication modules OMDi, OMDs, optical connectors CNc, CNo and optical fibers OF.
FIG. 1B shows the constitution of the optical communication module. The optical communication module OMD is constituted of an optical element block OBK including a laser diode LD and a photodiode PD, an analogue front end circuit AFE including a laser diode driver LDD and a transimpedance amplifier TIA, and an electric interface block SDC connected with an LSI. To explain the transmission operation of the optical communication module OMD, the laser driver LDD which receives an electrical signal from the electric interface block SDC drives the laser diode LD in response to a current signal so that an optical signal is outputted to an optical fiber OFtx from the laser diode LD.
On the other hand, to explain the receiver operation of the optical communication module OMD, an optical signal from an optical fiber OFrx is converted into a current signal by a photodiode PD, and the current signal is converted into a voltage signal by the transimpedance amplifier TIA and the voltage signal is transmitted to the electric interface block SDC. To suppress electric power consumed by a series of optical communication operations, it is important to suppress optical signal energy to be transmitted or received to a small value, and it is also important to suppress electric power consumed by the optical element block OBK and the analogue front end circuit AFE.
The suppression of the optical signal energy to be transmitted or received or electric power consumed by the laser LD to a small value leads to the suppression of energy of the optical signal which the photodiode PD receives so that an input current signal of the transimpedance amplifier TIA can be reduced. In general, the input current signal of the transimpedance amplifier TIA is a minute current of not more than several 100 μA. In future, to cope with the further lowering of electric power for in-device optical transmission, it is necessary to further lower the level of a receivable input current signal.
As a gain GTIA of the transimpedance amplifier TIA, approximately 1KΩ is necessary for ensuring several 100 mV necessary for operating the electric interface block SDC of a succeeding stage as an output voltage signal. The transimpedance amplifier TIA converts an electric current into a voltage with amplification and hence, a unit of a gain becomes Ω (Vo=GTIA*Iin).
FIG. 2A shows the block constitution of the transimpedance amplifier TIA described in patent literature 1. The transimpedance amplifier TIA includes a pre-amplifier PRAMP which converts a single-phase current signal from the photodiode PD into a voltage signal, a post-amplifier PSAMP which differentiates and amplifies a single-phase output signal from the pre-amplifier PRAMP, and an error amplifier ERAMP and an auto-threshold voltage control circuit ATC1 which detect a signal center level (signal threshold voltage) based on an output signal from the post-amplifier PSAMP. A gain of the pre-amplifier PRAMP is G1, a gain of the post-amplifier PSAMP is G2, and the gain GTIA of the whole transimpedance amplifier TIA is G1×G2 (GTIA=G1×G2).
The error amplifier ERAMP includes an operational amplifier OPAMP and resistors R1, R2, and a gain G3 of the error amplifier ERAMP is R2/R1 (G3=R2/R1). The auto-threshold voltage control circuit ATC1 is a primary low pass filter having a resistor R3 and a capacitor C3, and cut-off frequency f1 of the auto-threshold voltage control circuit ATC is 1/(2π×R3×C3 (f1=1/(2π×R3×C3).
Although it is necessary to increase the gain G1 of the initial-stage pre-amplifier PRAMP to enhance light receiver sensitivity of the transimpedance amplifier TIA, the gain G1 and an operational bandwidth have a tradeoff relationship. Accordingly, for example, to set the operational bandwidth to several GHz or more, the gain G1 of the pre-amplifier PRAMP must be held only at a level of approximately several 100Ω.
As described previously, the input current signal of the transimpedance amplifier TIA is approximately several 100 μA and hence, the output signal from the pre-amplifier PRAMP becomes approximately several 10 mV. The post-amplifier PSAMP includes a differential circuit in general and hence, an offset voltage Voff of approximately several 10 mV is generated at an input of the post-amplifier PSAMP due to irregularities in a pair transistor or load resistance. Particularly, when a field effect transistor (hereinafter abbreviated as MOS transistor) which is effectively used for high-speed communication and low power consumption is used, a value of the offset voltage Voff is large and hence, the output signal of the pre-amplifier PRAMP has substantially the same voltage as the offset voltage Voff whereby a high-quality current/voltage signal conversion operation is impaired by the offset voltage Voff.
To compensate for this offset voltage Voff, in patent literature 1, a negative feedback path is formed of the post-amplifier PSAMP, the error amplifier ERAMP and the auto-threshold voltage control circuit ATC1. An open loop gain Gopen of the negative feedback path is G2×G3 (Gopen=G2×G3). Due to the provision of the negative feedback path, the offset voltage Voff is reduced by an amount corresponding to the open loop gain Gopen so that the offset voltage Voff′ is Voff/Gopen (Voff′=Voff/Gopen).
FIG. 2B shows frequency characteristics of an open loop of the negative feedback path and the transimpedance amplifier TIA. In FIG. 2B, the open loop gain Gopen of the feedback circuit is taken on a left axis of ordinates, and the gain GTIA of the transimpedance amplifier TIA is taken on a right axis of ordinates. Although the open loop gain Gopen becomes constant at G2×G3 in a low frequency region, when the frequency exceeds f1, the open loop gain Gopen is decreased with inclination of 20 dB/dec due to a characteristic of the low pass filter of the auto-threshold voltage control circuit ATC1. On the other hand, although the gain GTIA of the transimpedance amplifier TIA is constant at G1×G2 in a high frequency region, a unity gain frequency f2 at which the gain of the negative feedback path becomes 1 is f1×G2×G3 (f2=f1×G2×G3) and hence, the gain GTIA is decreased with the inclination of 20 dB/dec in a region where the frequency is not more than f2.
Accordingly, the low cutoff frequency of the transimpedance amplifier TIA is determined based on the unity gain frequency f2 of the negative feedback path. To realize the low power consumption and the reduction of noises of the transimpedance amplifier TIA, it is necessary to decrease the number of stages of amplifiers which constitute the post-amplifier PSAMP by increasing the gain G1 of the pre-amplifier PRAMP and by lowering the gain G2 necessary for the post-amplifier PSAMP.
Accordingly, the gain G2 of the post-amplifier PSAMP is approximately 10 times, and it is necessary to increase the gain G3 of the error amplifier ERAMP to G3′ to enhance the compensation accuracy of the offset voltage Voff. However, when the gain of the error amplifier ERAMP is increased, a unity gain frequency f3 of the negative feedback path is also increased to f1×G2×G3′ (f3=f1×G2×G3′). As a result, the low cutoff frequency of the transimpedance amplifier TIA is increased to f3 from f2.
In general, the low cutoff frequency of the transimpedance amplifier TIA is required to be not more than 100 KHz. The capacitor C3 which can be mounted in a chip is approximately 100 pF due to the restriction imposed on an area which can be formed on a semiconductor and hence, the low pass filter characteristic of the auto-threshold voltage control circuit ATC1 is limited to approximately 100 kHz. Accordingly, due to the restriction imposed on required lower-region cutoff frequency of the transimpedance amplifier TIA, it is difficult to enhance the compensation accuracy of the offset voltage Voff with the increase of the gain G3 of the error amplifier ERAMP, and the offset voltage Voff becomes approximately 1/10. Accordingly, with the output of the pre-amplifier PRAMP, the offset voltage Voff of several mV is generated with respect to a receiver signal of several 10 mV and hence, a rate of the offset voltage Voff with respect to the receiver signal exceeds several 10% whereby it is difficult to realize the high-quality and high-speed receiver operation of several Gbsp or more.